1. Field of the Invention
The present invention is concerned with wireless communications apparatus, and associated methods, operable to make wireless communication emissions on the basis of the block transmission approach.
2. Discussion of Background
Aspects of the invention can apply to any wireless communication devices that use block transmissions when channel state information is available at the transmitter. Examples of block transmissions are orthogonal frequency division multiplexing (OFDM) and single carrier block transmission with frequency domain equalization (SC-FDE).
Aspects of the invention may apply to any products that use OFDM and/or SC-FDE transmissions. Examples of these include devices compliant with IEEE 802.11 a/b/g/n, IEEE 802.16 e/j/m, and potentially IEEE 802.15.3c 60 GHz devices.
The field of the invention will now be described, with reference to a description of conventional wireless communications technology employing block transmission with cyclic prefix insertion and removal.
Block transmission for use in wireless communications has been widely investigated in published literature (e.g., Z. Wang and G. B. Giannakis, “Wireless Multicarrier Communications: Where Fourier Meets Shannon” IEEE Signal Processing Magazine, Vol. 17, No. 3, pp. 29-48, May 2000). Two well-known block transmission schemes are Orthogonal frequency-division multiplexing (OFDM) and Single carrier with frequency domain equalization (SC-FDE), which have been widely adopted in world-wide standards such as IEEE 802.11, IEEE 802.16 and IEEE 802.15.3c.
Block diagrams of the conventional OFDM and SC-FDE systems with a single transmit and receive antenna are shown in FIGS. 1 and 2 respectively.
FIG. 1 illustrates an example of a communication system 10 in schematic form. A transmit train 12 is illustrated receiving a bit sequence, which is input to a convolutional encoder 20 and then to an interleaver 22. Symbol mapping 24 is then applied to the data, which is then converted from serial to parallel 26. The parallel data is then converted from frequency to time domain by an IFFT 28. A cyclic prefix is then inserted 30 and then, prior to transmission on a single antenna, the data is converted back from parallel to serial 32.
A transmission channel is illustrated as represented by a channel matrix H. A receiver train 14 as illustrated receives transmitted signals on a single antenna, and these transmitted signals are converted from serial to parallel 40 and then the cyclic prefix is removed 42. Thereafter, a fast Fourier converts the signal from time to frequency domain and the resultant frequency subcarriers are converted from parallel to serial. Symbol de-mapping takes place 48 and then the de-mapped symbols are deinterleaved 50 and a Viterbi decoder 52 extracts a bit sequence.
FIG. 2 illustrates a similar arrangement of transmitter train, channel and receiver train, for SC-FDE. In that arrangement, the IFFT 28 of the transmitter train in FIG. 1 is omitted, and, between the FFT 44 and the parallel to serial converter 46 of the receiver train, an equaliser 45 (which can be a frequency domain zero-forcing equalizer or a frequency domain minimum mean-squared error equalizer) and an IFFT 47 are included. This takes account of the different operation of SC-FDE over OFDM.
In both systems, a cyclic prefix (CP) of length L, which is a copy of the last L symbols in a block, is inserted at the beginning of this block at the transmitter. At the receiver, the first L received symbols in each block are discarded. The insertion and removal of the CP is illustrated in FIGS. 3 and 4, where x (FIG. 3) is a block of transmitted symbols before CP insertion, and y (FIG. 4) is the block of the received data before CP removal.
When the length of the CP is greater than or equal to the channel delay spread, the insertion and removal of CP at the transmitter and the receiver converts the Toeplitz channel matrix H into a circulant one ({tilde over (H)}), which can be decomposed as {tilde over (H)}=FHΛF, where F denotes the Fourier transform matrix, FH denotes its Hermitian transpose, and Λ is a diagonal matrix with its kth diagonal entry being the kth frequency domain channel coefficient.
With the subsequent fast Fourier transform (FFT) 44, the resulting channel can be considered as flat over each OFDM subcarrier for an OFDM system, or over each frequency domain symbol of the transmitted block for an SC-FDE system, thus allowing a simple, one-tap frequency domain equalizer to remove the inter-symbol interference in both systems.
It should be noted that, for systems with channel coding as shown in FIG. 1, the equalizer may be included in the Viterbi soft decoder to improve the performance.
The examples given in FIGS. 1 and 2 are concerned with baseband transmission. In the discrete time domain, each channel tap is a complex number. The frequency domain channel coefficients, which contain the Fourier transform of the time domain channel taps, are complex as well.
Various methods of generating a circulant channel matrix exist. For example, zero padding (ZP) or cyclic suffix (CS) can also be used. The use of CP and/or CS in OFDM for the purpose of precise symbol timing is detailed in J. Chun, B. Ihm, and Y. Jin, “Method for detecting OFDM symbol timing in OFDM system”, (LG Electronics Inc., International Patent WO 2006/019255, February 2006). However, in that publication, no receiver structure is discussed.
Time reversal (TR) is a feedback technique that has attracted much interest in acoustic and medical applications. Recently, the application of TR to wireless communication has also been investigated. In a TR system, channel state information is made available at the transmitter, through feedback from the receiver (if frequency-division duplexing is employed) or by utilising the reciprocity of the channel (if time-division duplexing is employed).
For example, in a system in which a wireless channel has a channel impulse response h(t), a TR system prefilters the transmitted signal by the time-reversed, complex-conjugated channel impulse response, and then transmits the resulting signal.
A TR system with M transmit antennas and one receiver antenna is illustrated in FIG. 5, where h*k(−t) is the prefilter applied on the k th transmitted antenna. The prefilter is the time-reversed, complex-conjugate of hk(t), which is the channel impulse response between the kth transmitted antenna and the receiver.
Due to prefiltering, the equivalent composite channel, which is the convolution of the TR prefilter and the channel impulse response, is essentially the auto-correlation of the channel impulse response, i.e.,g(t)=h(t)h*(−t)where  denotes convolution. The equivalent channel impulse response, therefore, has an enhanced energy at and near its centre, and a diminished energy at the tail. This is the “time focusing” property of a TR system. In a multi-antenna/multiple access scenario, TR also provides “space focusing” where energy can be focused to an intended receive antenna or an intended user.
The “time focusing” and “space focusing” properties of TR have been exploited in communication systems. For example, the “time focusing” property essentially allows for fewer time domain equalizer taps at the receiver. Examples can be found in impulse radio ultra-wideband (UWB) systems. Unfortunately, the study of TR was limited to symbol-by-symbol transmissions, and few (if any) researchers have considered the complex symmetry of the resulting equivalent channel taps due to the TR prefilter.
Space-time block coding (STBC) has received considerable interest in systems with multiple transmit antennas. The advantage of orthogonal space-time block coding (OSTBC) is the simplicity of the decoder. One well-known example of OSTBC is Alamouti space-time block coding for two transmit antennas, which achieves full rate (rate one) and full diversity for both real and complex signalling.
Using the theory of orthogonal design, the Alamouti STBC can be extended for more than two transmit antennas using a generalized orthogonal design. Under certain circumstances and for systems with M transmit antennas, a generalized orthogonal matrix with size p×M and entries 0, ±x1, ±x*1, . . . , ±x*k, ±xk can be formed, and the transmission rate is defined as R=k/p.
It has been shown that, when complex signalling is transmitted (e.g., QPSK), full rate (R=1) orthogonal designs exist only for two transmit antennas. However, full rate OSTBC exists for M≧2 transmit antennas when real constellations such as BPSK are used.
For systems transmitting complex signalling using more than two transmit antennas, previous attempts have focused on seeking orthogonal designs at the expense of a rate loss (e.g., rate ¾ and rate ½ OSTBC), or using quasi-orthogonal designs with full rate at the expense of increasing decoder complexity or a losing diversity.
Utilizing channel state information at the transmitter (CSIT), transmit precoding has been extensively investigated for systems with multiple transmit antennas, to increase either transmission rate, coverage, or robustness, and has also been considered in future wireless applications (e.g., IEEE 802.16).
The transmitter structure with precoding is illustrated in FIG. 6, where W is the precoding matrix/vector, and C is the codeword encoded from the transmitted signal. The codeword can be a space-time block coded signal of size M×T, with M being the number of transmit antennas and T being the time slots, or simply a scalar transmitting on the kth transmitter antenna, weighted by the kth entry of the precoding vector, wk. It has been shown that the optimal transmission, that minimizes the average pairwise codeword error probability (PEP), is the single mode eigen-beamforming, where the precoding vector is the dominant eigenvectors of HtHHt, where Ht is the MISO channel matrix of size 133 M. In block transmissions such as OFDM, precoding may be applied on a per subcarrier basis.
The present description of the field of the invention will now consider, by way of example, a system with multiple transmit antennas and one receive antenna (i.e., multiple input and single output (MISO)), with perfect CSIT.
Prior art examples will now be used to illustrate issues apparent in the field of the invention.
When a channel impulse response is real-valued, convolution of the channel impulse response with its time-reversed counterpart provides symmetry (not only complex symmetry) in the resulting equivalent channel. Exploiting this symmetry, a discrete cosine transform (DCT) can be used instead of an FFT in a multicarrier system to diagonalise the channel matrix, resulting in real-valued DCT channel coefficients. This method was proposed and discussed in N. Al-Dhahir, H. Mimi and S. Satish, “Optimum DCT-Based Multicarrier Transceivers for Frequency-Selective Channels”, IEEE Trans. on Comm. Vol. 54, May 2006, pp. 911-921. The main differences between Al-Dhahir and the conventional multicarrier systems are that:    1. A front-end prefilter at the receiver is used in Al-Dhahir. The prefilter coefficients are designed asw=Ryy−1Ryxgwhere Ryx and Ryy are the output-input cross correlation and the output autocorrelation matrices, respectively, and g is the equivalent channel obtained by convolving the original channel impulse response and its time reversed channel. Alternatively, as discussed in Al-Dhahir, the prefilter coefficients can also be designed as the time-reversed channel, resulting in an overall symmetric equivalent channel impulse response. It will be observed that the abovementioned symmetry of an equivalent channel can only be obtained when the channel impulse response is real-valued, which is usually not true when baseband transmission is considered and when complex signalling is transmitted.    2. When the equivalent channel impulse response is symmetric, the conventional IDFT and DFT used at the transmitter and receiver in a multicarrier system can be replaced by the IDCT and DCT operations, resulting in real-valued DCT channel coefficients. A transceiver block diagram of the DCT-based multicarrier system for real signalling is shown in FIG. 7. When the transmitted signal is complex-valued, the equivalent channel becomes complex-symmetric, the above illustrated transceiver structure has to be tailored into two branches to process the real and complex components of the complex signals respectively, as is illustrated in Al-Dhahir. A transceiver block diagram of the DCT-based multicarrier approach to complex signalling is shown in FIG. 8.
The use of TR in STBC was presented in E. Lindskog and A. J. Paulraj, “Time-reversal block transmit diversity system for channels with intersymbol interference and method”, (Standford, International Patent WO/2001/080446, October 2001), in which transmitted signals (rather than the channel impulse responses) are time reversed, and complex conjugated before transmitting from one of the given transmit antennas in one of the given time slots. The transmission block format of Lindskog and Paulraj is presented in FIG. 9, where “GS” denotes guard sequence and conventional CP insertion is used where the last L data symbols are “copied” to the beginning of each block.
In Lindskog and Paulraj, expressing the length-L discrete frequency-selective channel h=[h1, h2, . . . , hL−1]T by a polynomial in the unit delay operator q−1 ash(q−1)=h0+h1q−1+ . . . +hL−1q−L+1 the noiseless received signal can be written asy(t)=h(q−1)x(t)where x(t) is the symbol sequence transmitted from the antenna. Or, in the discrete form, the received signal can be expressed asy(q−1)=h(q−1)x(q−1).
Considering two transmit antennas with the transmission format given by FIG. 9, the noiseless received signals on the 1st and the 2nd antenna are given byz1=y(k)(q−1)=h1(q−1)x1(q−1)+h2(q−1)x2(q−1)andz2=(y(k+1))*(q)=h*2(q)x1(q−1)−h*1(q)x2(q−1).
Expressing z1 and z2 in a matrix form:z=Hxwhere z=[z1,z2]T, x=[x1,x2]T, and
      H    ⁡          (              q                  -          1                    )        =            [                                                                  h                1                            ⁡                              (                                  q                                      -                    1                                                  )                                                                                        h                2                            ⁡                              (                                  q                                      -                    1                                                  )                                                                                                        h                2                *                            ⁡                              (                q                )                                                                        -                                                h                  1                  *                                ⁡                                  (                  q                  )                                                                        ]        .  
After 2-D matched filtering at the receiver, the received signal becomes
                              [                                                                      r                  1                                                                                                      r                  2                                                              ]                =                                            H              H                        ⁡                          (                              q                                  -                  1                                            )                                ⁢                      H            ⁡                          (                              q                                  -                  1                                            )                                ⁢          x                                        =                              [                                                                                                      h                      eq                                        ⁡                                          (                                              q                                                  -                          1                                                                    )                                                                                        0                                                                              0                                                                      -                                                                  h                        eq                                            ⁡                                              (                                                  q                                                      -                            1                                                                          )                                                                                                                  ]                    ⁢          x                    whereheq(q−1)=h*1(q)h1(q−1)+h*2(q)h*2(q−1).
The detection of x1 and x2 thus decouples. It will be noted that the inter-symbol interference (ISI) still exists in each block. This is resolved in Lindskog and Paulraj by using the maximum likelihood sequence estimation (MLSE) algorithm.
In addition to TR-STBC, both OFDM-STBC and SCFDE-STBC have also been described such as in D. Wang, L. Jiang, and C. He, “A MIMO Transceiver Scheme using TR-STBC for Single-Carrier UWB Communications with Frequency domain Equalization”, (IEEE CHINACOM'07, August 2007, pp. 1142-1146).
Although real-valued DCT channel coefficients can be generated using the method provided in the first prior art reference Al-Dhahir, this is limited to real signalling only. When complex signals are transmitted, the convolution of a time reversed prefilter at the receiver:
“results in a conjugate-symmetric overall CIR, where only the real part is symmetric, whereas the imaginary part will be antisymmetric, and hence not diagonalizable by the type-II DCT” Al-Dhahir.
Therefore, when complex signals such as QPSK are used, real-valued DCT channel coefficients cannot be obtained, and two branches of IDCT/DCT have to be used to deal with the real and complex components of the complex signal, indicating an increase in complexity.
There are two major problems in the second prior art document Lindskog and Paulraj:    1. As for the orthogonal designs of STBC code for complex constellation, the arrangement described in Lindskog and Paulraj is limited to two transmit antennas when full rate transmission is desired. The decoupling of two blocks is based on the orthogonality of the MIMO channel matrix H(q−1), which can be realised by an Alamouti-like transmission structure for two transmit antennas. When the number of transmit antennas is greater than two, full rate orthogonal designs with rate one do not exist for complex constellations, and therefore cannot be used to achieve full rate by directly using the method provided in Lindskog and Paulraj.    2. Inter-symbol interference exists in each block, and the use of MLSE for signal detection in the time domain can be computational complex.
As with TR-STBC (Lindskog and Paulraj) other prior art technologies exist that apply STBC to block transmissions, where a block of symbols that are either time-reversed and complex-conjugated or simply complex-conjugated before transmitting over one of the given antennas in one of the given time slots. These prior art technologies are considered as extensions of STBC to block transmissions. Examples are TR-STBC, OFDM-STBC, and SCFDE-STBC (Wang and Jiang).
In these extensions, the CP insertion and removal follows that of conventional block transmissions (e.g., FIGS. 3 and 4), and the frequency domain channel coefficients are complex-valued. This limits the use of full rate OSTBC to systems with more than two transmit antennas when complex constellations are employed, since full rate OSTBC only exists for real constellations when the number of transmit antennas is greater than two. As a simple example, when the number of transmit antennas is four, and when symbols are mapped onto a complex constellation such as QPSK, none of the prior art technologies can achieve full rate using the existing OSTBC.
None of the abovementioned prior art examples consider the complex-symmetry of the equivalent channel in a TR system. Although the channel symmetry of a TR system was explored in Al-Dbahir, the real-valued DCT channel coefficients are unfortunately limited to real signalling only, and two branches of DCTs are required to process the real and imaginary components of a signal when it is complex-valued.